Switching-mode power supply having a synchronous rectifier

ABSTRACT

A DC-to-DC converter incorporates a transformer having a primary winding connected to a pair of DC input terminals via an active switch, which turns on and off under the control of a feedback circuit, and a secondary winding connected to a pair of DC output terminals via a synchronous rectifier and a smoothing capacitor. The synchronous rectifier is a parallel connection of a synchronous rectifier switch and a diode. A synchronous rectifier control circuit is connected to the synchronous rectifier switch for causing conduction therethrough while the active switch is off. The synchronous rectifier control circuit comprises a capacitor for determination of the conducting periods of the synchronous rectifier switch, and a logic network for on/off control of the synchronous rectifier switch according to whether the active switch is on or off and whether the capacitor voltage is higher than a predefined threshold or not.

CROSS REFERNCE TO RELATED APPLICATIONS

This is a continuation of Application PCT/JP2004/014075, filed Sep. 27,2004, which claims priority to Japanese Patent Application No.2003-340758 filed Sep. 30, 2003.

BACKGROUND OF THE INVENTION

This invention relates to an electronic power supply unit known as theswitching-mode power supply, incorporating a switching regulator wherebythe load current is switched rapidly on and off for output voltagestabilization. More specifically, the invention pertains to such a powersupply of the kind having a synchronous rectifier circuit.

A typical conventional switching-mode power supply with a flybackDC-to-DC converter includes a transformer having a primary windingcoupled to the pair of DC input terminals via an active switch, and asecondary winding coupled to the pair of DC output terminals via arectifying diode and smoothing capacitor. The DC input voltage is turnedon and off as the active switch is driven by pulses that have beenwidth-modulated by a feedback circuit monitoring the DC output voltage.Energy is stored on the transformer during the conducting periods of theactive switch and released during its nonconducting periods. Thesmoothing capacitor is charged as the rectifying diode conducts duringthe nonconducting periods of the active switch.

A voltage drop of approximately 0.8 volt has been known to occur acrossthe rectifying diode of the above switching-mode power supply. JapaneseUnexamined Patent Publication No. 9-163736 teaches how to reduce thisvoltage drop, and consequent power loss, across the rectifying diode.Connected in parallel with the rectifying diode according to this priorart is a synchronous rectifier switch which is turned on during theconducting periods of the rectifying diode. The synchronous rectifierswitch, particularly when in the form of a bipolar or field-effecttransistor, introduces a voltage drop of as low as 0.2 volt or so,realizing an appreciable diminution of an overall voltage drop on theoutput side of the transformer.

This prior art technique has proved to possess its own shortcomings,however. One of them is the difficulty of turning on the synchronousrectifier switch in exact synchronism with the conducting periods of therectifying diode. This difficulty becomes even more serious because theconducting periods of the rectifying diode are subject to change withthe input voltage and with the voltage requirement of the load.

Another weakness has manifested itself in the event of an abrupt drop inthe output voltage of the switching-mode power supply in response to theload. Thereupon the standard feedback circuit of the closed-loopswitching regulator has responded by correspondingly extending theconduction time of the active switch. The possible result has been theoverlapping of the conduction periods of the active switch and those ofthe synchronous rectifier switch. Such overlapping, if it occurred atall, led to noise production and, worse yet, to the destruction of theassociated circuit parts.

The synchronous rectification technology and its yet-unremedieddrawbacks discussed above are not peculiar to the flyback DC-to-DCconverter. The same scheme is applicable, at the risk of the emergenceof like difficulties, to many other varieties of switching-mode powersupplies, some known examples being a boost converter, a forwardconverter, a chopper controller, and a combination of an inverter and arectifier/filter circuit.

SUMMARY OF THE INVENTION

The present invention seeks to resolve the noted problems attendant tothe synchronous rectifier switch in switching-mode power supplies ofvarious known types and configurations.

Briefly, the invention may be summarized as a switching-mode powersupply for DC-to-DC conversion, comprising a converter circuit connectedbetween DC input means and DC output means. The converter circuitincludes an active switch for switching the input DC voltage on and offunder the direction of a switch control circuit. Connected between theconverter circuit and the DC output means is a synchronous rectifierhaving a parallel connection of a diode and a switch. For on/off controlof this synchronous rectifier switch there is provided a synchronousrectifier control circuit comprising: (a) conduction period detect meansfor providing a conduction period detect signal indicative of whetherthe active switch is conductive or not; (b) a capacitor fordetermination of the conducting periods of the synchronous rectifierswitch; (c) a charge/discharge circuit connected to the convertercircuit and the capacitor for causing the latter to be charged anddischarged according to whether the active switch is conductive ornonconductive and hence for causing the capacitor to develop a voltageindicative of the conduction or nonconduction of the active switch; and(d) a logic network having an input connected to the conduction perioddetect means, another input connected to the capacitor, and an outputconnected to the synchronous rectifier switch, for making on/off controlof the synchronous rectifier switch according to whether the activeswitch is conducting or nonconducting and whether the voltage across thecapacitor is higher than a predefined voltage or threshold or not.

The term “synchronous rectification” as used herein and in the claimsappended hereto refers to all sorts of rectifications of the converteroutput in certain phase relationships to the conduction andnonconduction of the active switch. The term “synchronous rectifierswitch” likewise refers to a switch conducive to the rectification orsmoothing of the converter output in certain phase relationships to theconduction and nonconduction of the active switch.

A most pronounced feature of the invention resides in the configurationof the synchronous rectifier control circuit for on/off control of thesynchronous rectifier switch. Despite the simple design comprising acapacitor and a logic network, the synchronous rectifier control circuitaccurately determines the conducting periods of the synchronousrectifier switch and make them as long as feasible within the limits ofthe nonconducting periods of the active switch. A higher efficiency isthus gained by the switching-mode power supply. Furthermore, as theactive switch and synchronous rectifier switch are positively preventedfrom concurrent conduction, the power supply is saved from noiseproblems and the rupture of its constituent parts.

The above and other objects, features and advantages of this inventionwill become more apparent, and the invention itself will best beunderstood, from a study of the following description and appendedclaims, with reference had to the attached drawings showing somepreferable embodiments of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic electrical diagram of the switching-mode powersupply embodying the principles of this invention.

FIG. 2, consisting of (A) through (F), is a diagram showing waveformsappearing at various parts of FIG. 1 in proper time relationship to oneanother.

FIG. 3 is a schematic electrical diagram of another preferred form ofswitching-mode power supply embodying the invention.

FIG. 4 is a schematic electrical diagram of still another preferred formof switching-mode power supply embodying the invention.

FIG. 5 is a schematic electrical diagram of yet another preferred formof switching-mode power supply embodying the invention.

FIG. 6, consisting of (A) through (F), is a diagram showing waveformsappearing at various parts of FIG. 5 in proper time relationship to oneanother.

FIG. 7 is a schematic electrical diagram of a further preferred form ofswitching-mode power supply embodying the invention.

FIG. 8, consisting of (A) through (F), is a diagram showing waveformsappearing at various parts of FIG. 7 in proper time relationship to oneanother.

FIG. 9 is a schematic electrical diagram of a wave-shaping circuit foruse in the FIG. 1 embodiment in place of the NOT circuit of itssynchronous rectifier control circuit.

FIG. 10 is a partial schematic electrical diagram of a further preferredform of switching-mode power supply embodying the invention.

FIG. 11 is a schematic electrical diagram of a further preferred form ofswitching-mode power supply embodying the invention.

FIG. 12, consisting of (A) through (D), is a diagram showing waveformsappearing at various parts of FIG. 11 in proper time relationship to oneanother.

FIG. 13 is a schematic electrical diagram of a modification of thesynchronous rectifier control circuit in the FIG. 1 embodiment.

FIG. 14 is a schematic electrical diagram of another modification of thesynchronous rectifier control circuit in the FIG. 1 embodiment.

FIG. 15 is a schematic electrical diagram of a further modification ofthe synchronous rectifier control circuit in the FIG. 1 embodiment.

FIG. 16 is a schematic electrical diagram of a further modification ofthe synchronous rectifier control circuit in the FIG. 1 embodiment.

FIG. 17 is a schematic electrical diagram of a further modification ofthe synchronous rectifier control circuit in the FIG. 1 embodiment.

FIG. 18 is a schematic electrical diagram of a further preferred form ofswitching-mode power supply embodying the invention.

FIG. 19 is a schematic electrical diagram of a further preferred form ofswitching-mode power supply embodying the invention.

FIG. 20 is a schematic electrical diagram of a further preferred form ofswitching-mode power supply embodying the invention.

FIG. 21 is a schematic electrical diagram of a further preferred form ofswitching-mode power supply embodying the invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The present invention is currently believed to be best embodied in theswitching-mode power supply of flyback DC-to-DC converter type diagramedin FIG. 1 of the above drawings. Broadly, the exemplified flybackDC-to-DC power supply comprises:

1. A pair of DC input terminals 1 _(a) and 1 _(b) as the DC input means.

2. A converter circuit 2 including an active switch Q₁ for switching theDC input on and off.

3. A synchronous rectifier circuit 3, to which the present inventionparticularly pertains, connected to the outputs of the converter circuit2.

4. A pair of DC output terminals 4 _(a) and 4 _(b) as the DC outputmeans connected to the converter circuit 2 via the synchronous rectifiercircuit 3 and shown connected to a load 4 to be powered.

5. A switch control circuit 5 connected between the pair of DC outputterminals 4 _(a) and 4 _(b) and the converter circuit 2 for controllablydriving the active switch Q₁.

6. A smoothing capacitor C_(o) connected between the pair of DC outputterminals 4 _(a) and 4 _(b), although this capacitor in combination withthe DC output terminals 4 _(a) and 4 _(b) might be collectively calledDC output means.

The pair of DC input terminals 1 _(a) and 1 _(b) are shown connected toa source 1 of DC voltage. The DC voltage source 1 may be either abattery or a rectifier/filter circuit connected to a commercial sourceof AC power.

The converter circuit 2 includes a transformer 6 in addition to theactive switch Q₁. The transformer 6 has a primary winding N₁ andsecondary winding N₂, both coiled around a magnetic core 6′ andelectromagnetically coupled together. As indicated by the dots in FIG.1, the transformer windings N₁ and N₂ are oppositely polarized withrespect to each other in this particular embodiment. It is understoodthat the transformer 6 is additionally furnished with a tertiarywinding, not shown, for powering the switch control circuit 5.

Shown as an insulated-gate field-effect transistor (IGFET), the activeswitch Q₁ has a first main electrode or drain connected to the DC inputterminal 1 _(a) via the transformer primary N₁, a second main electrodeor source connected to the other input terminal _(b), which is grounded,and a control electrode or gate connected to the switch control circuit5.

The transformer secondary N₂ has its opposite extremities coupled to thepair of DC output terminals 4 _(a) and 4 _(b) via the synchronousrectifier circuit 3 and smoothing capacitor C_(o). The voltage acrossthe transformer secondary N₂ is therefore rectified by the synchronousrectifier circuit 3 and smoothed by the smoothing capacitor C_(o).

The synchronous rectifier circuit 3 is broadly divisible into asynchronous rectifier Q₂ and a synchronous rectifier control circuit 7.The synchronous rectifier Q₂ is also shown as an IGFET compounded of aswitch 8 and a diode D₀ which are connected in parallel therewith.Constituting the major part of the IGFET, the synchronous rectifierswitch 8 has a drain connected to the transformer secondary N₂, a sourceconnected to the negative output terminal 4 _(b), and a gate connectedto the synchronous rectifier control circuit 7. The synchronousrectifier switch 8 is therefore in parallel with the serial circuit ofthe transformer secondary N₂ and smoothing capacitor C₀. The diode D₀ isbuilt into the synchronous rectifier Q₂ by being formed on the samesubstrate, as of silicon, as the synchronous rectifier switch 8. Thisdiode could, however, be a discrete part electrically connected inparallel with the synchronous rectifier switch.

In the case where the synchronous rectifier Q₂ is of a siliconsemiconductor, the voltage drop across the synchronous rectifier switch8 when it is closed is approximately 0.2 volt. The voltage drop acrossthe conducting diode D₀ on the other hand is approximately 0.8 volt. Aless voltage drop and less power loss will occur if the voltage acrossthe transformer secondary N₂ is rectified while the synchronousrectifier switch 8 is held conducting, than if only the diode D₀ isconducting.

As the name implies, the synchronous rectifier control circuit 7 of thesynchronous rectifier circuit 3 makes on/off control of the synchronousrectifier switch 8. It comprises:

1. A capacitor C₁ for providing a capacitor voltage V_(c1), with awaveform diagramed at (C) in FIG. 2, indicative of whether thesynchronous rectifier switch 8 may, and may not, be held conductive.

2. A charge/discharge circuit 9 through which the capacitor C₁ is to becharged and discharged according to whether the active switch isconductive or not.

3. A conduction period detector circuit 11 for ascertaining whether theactive switch Q₁ is conductive or not.

4. A logic network 20 through which both capacitor C₁ and conductionperiod detector circuit 11 are connected to the gate of the synchronousrectifier switch 8 for causing conduction therethrough in relation tothe conduction and nonconduction of the active switch Q₁.

It is understood that the capacitor C₁ of the synchronous rectifiercontrol circuit 7 is less in capacitance than the smoothing capacitorC₀. The charge/discharge circuit 9 comprises a serial connection ofdiode D₁ and resistor R₁ for charging the capacitor C₁, and anotherresistor R₂ for discharging the same. The serial circuit of diode D₁ andresistor R₁ is connected to the drain D of the synchronous rectifier Q₂by way of a conductor 21 on one hand and, on the other, to one terminalof the capacitor C₁. The other terminal of this capacitor C₁ isconnected both to the source S of the synchronous rectifier Q₂ and tothe DC output terminal 4 _(b) by way of a conductor 22. The diode D₁ isso oriented as to be forward biased by the voltage induced across thetransformer secondary N₂ during the conducting periods of the activeswitch Q₁. The resistor R₂ is connected in parallel with the capacitorC₁. The illustrated configuration of the charge/discharge circuit 9 issubject to a variety of modifications, as will be later referred to inmore detail.

The conduction period detector circuit 11 of the synchronous rectifiercontrol circuit 7 is a combination of two voltage-dividing resistors R₃and R₄ interconnected in series and themselves connected between thedrain D and source S of the synchronous rectifier Q₂. Thus the potentialat the junction 13 between these voltage-dividing resistors R₃ and R₄represents a predefined fraction of the voltage between the drain andsource of the synchronous rectifier Q₂.

The potential at the junction 13 between the voltage-dividing resistorsR₃ and R₄ switches between two values depending upon whether the activeswitch Q₁ is conducting or not. During the conducting periods of theactive switch Q₁ the sum of the voltage across the transformer secondaryN₂ and that across the smoothing capacitor C₀ is applied to thesynchronous rectifier Q₂. The junction 13 between the voltage-dividingresistors R₃ and R₄ gains a high potential as that high voltage acrossthe synchronous rectifier Q₂ is divided by the resistors R₃ and R₄.During the nonconducting periods of the active switch Q₁, on the otherhand, the synchronous rectifier Q₂ conducts thereby causing the voltagebetween its drain and source to go low. The potential at thevoltage-dividing resistor junction 13 also goes low. Thus the conductionperiod detector circuit 11 provides a binary signal indicative of theconduction or nonconduction of the active switch Q₁, for delivery to thelogic network 20. The binary output from the conduction period detectorcircuit 11 is utilized by the logic network 20 for determination of thesignal for driving the synchronous rectifier switch 8.

The logic network 20 of the synchronous rectifier control circuit 7 hasa first input 26 connected to the conduction period detector circuit 11,a second input 27 connected to the capacitor C₁, and an output 28connected to the gate G of the synchronous rectifier switch 8. The logicnetwork 20 performs the following three functions for on/off control ofthe synchronous rectifier switch 8:

1. Comparison of the voltage V_(c1) across the capacitor C₁ with apredefined voltage or threshold.

2. Determination of the moment the synchronous rectifier switch 8 is tobe set out of conduction, on the basis of the moment the capacitorvoltage V_(c1) drops past the threshold.

3. Determination of the conducting periods of the synchronous rectifierswitch 8 on the bases of the result of comparison of the capacitorvoltage V_(c1) with the threshold and the binary output from theconduction period detector circuit 11.

More specifically, in this particular embodiment, the logic network 20comprises a NOT circuit 10 and NOR circuit 12. Connected to thecapacitor C₁, the NOT circuit 10 goes low when the voltage across thecapacitor is higher than the threshold, and high when the capacitorvoltage is less than the threshold. Thus the NOT circuit 10 functions toshape the infinitely varying capacitor voltage into a binary signal.

The above function of the NOT circuit 10 will be better understood froman inspection of (C) and (D) in FIG. 2. Shown at (C) in this figure isthe voltage V_(c1) across the capacitor C₁ and at (D) the output fromthe NOT circuit 10. The NOT circuit 10 has the threshold V_(th1). Theoutput from the NOT circuit 10 is high when the capacitor voltage V_(c1)is less than the threshold V_(th1), as from t₂ to t₄. The output fromthe NOT circuit 10 is low when the capacitor voltage V_(c1) is higherthan the threshold V_(th1), as from t₄ to t₆. Thus, constantly comparingthe continually varying input with the fixed value, the NOT circuit 10translates the capacitor voltage V_(c1) into the binary signalpreparatory to delivery to the NOR circuit 12.

The NOR circuit 12 has one input connected to the NOT circuit 10,another input to the junction 13 between the voltage-dividing resistorsR₃ and R₄ of the conduction period detector circuit 11, and an output tothe gate G of the synchronous rectifier switch 8. Thus, as will be seenfrom (E) in FIG. 2, the NOR circuit 12 causes conduction through thesynchronous rectifier switch 8, as from t₁ to t₂, when the inputs fromthe NOT circuit 10 and the conduction period detector circuit 11 areboth low. In other words, the synchronous rectifier switch 8 conductswhen the capacitor voltage V_(c1) is higher than the threshold V_(th1)and at the same time when the active switch Q₁ is off, designatedT_(off) at (A) in FIG. 2. It is understood that both NOT circuit 10 andNOR circuit 12 are energized from a DC power supply, not shown.

The smoothing capacitor C₀ is connected in parallel with the transformersecondary N₂ via the synchronous rectifier Q₂. The pair of DC outputterminals 4 _(a) and 4 _(b) are connected across the smoothing capacitorC₀. The synchronous rectifier Q₂ and smoothing capacitor C₀ constitutein combination the rectifying and smoothing circuit for the output fromthe converter circuit 2.

With continued reference to FIG. 1, inserted in the feedback path fromthe pair of DC output terminals 4 _(a) and 4 _(b) to the convertercircuit 2, the active switch control circuit 5 creates thepulse-width-modulated switch control signal V_(g), shown at (A) in FIG.2, thereby to drive the active switch Q₁ so as to keep the DC outputvoltage constant. The switch control circuit 5 includes an outputvoltage processing circuit 5 _(a) having inputs connected to the DCoutput terminals 4 _(a) and 4 _(b) by way of conductors 14 and 15,respectively, for providing an output voltage signal indicative of thevoltage between the pair of DC output terminals. The output voltagesignal is a voltage signal in direct proportion with the DC outputvoltage in this embodiment, although it could be in inverse proportionwith the DC output voltage. The output voltage processing circuit 5 _(a)is connected to the negative input of a comparator 5 _(c).

Also included in the switch control circuit 5 is a sawtooth generator 5_(b) which generates a sawtoothed voltage with a frequency of as high asfrom 20 kHz to 100 kHz. A triangular wave generator is a possible,obvious alternative to the sawtooth generator 5 _(b). The sawtoothgenerator 5 _(b) is connected to the positive input of the comparator 5_(c). The output from this comparator 5 _(c) is thepulse-width-modulated switch control signal V_(g), which is high whenthe sawtooth wave is higher than the output voltage signal. The switchcontrol signal V_(g) is delivered through the conductor 16 to the gateor equivalent control electrode of the active switch Q₁ in the convertercircuit 2. The active switch Q₁ is “on”, designated T_(on) at (A) inFIG. 2, when the switch control signal V_(g) is high, and off,designated T_(off), when the switch control signal is low.

The pulse-width-modulated switch control signal V_(g) is impressedbetween the gate and source of the active switch Q₁. The connectionbetween the comparator Sc and the source of the active switch Q₁ is notshown in FIG. 1.

Although shown connected to the pair of DC output terminals 4 _(a) and 4_(b) in FIG. 1, the switch control circuit 5 could be connected to theunshown tertiary winding of the transformer 2. Or, for that matter, itmight be connected to any parts having a voltage in proportion with theDC output voltage. Optionally, moreover, the feedback circuit maycontain an isolation mechanism such as the familiar opto-couplers toisolate it from the DC output.

Operation

The DC voltage from the source 1 is applied to the transformer primaryN₁ during each conducting period of the active switch 1. The transformer6 stores the energy because then the switch 8 and diode D₀ of thesynchronous rectifier Q₂ are both nonconductive. Assuming that thesmoothing capacitor C₀ has been charged, the sum of the voltage acrossthe transformer secondary N₂ and that across the smoothing capacitor C₀will cause conduction through the diode D₁ of the charge/dischargecircuit 9 during the conducting period T_(on) of the active switch Q₁,as from t₃ to t₅ in FIG. 2. The capacitor C₁ of the synchronousrectifier control circuit 7 will then be charged with a prescribed timeconstant, so that the voltage V_(c1) across the capacitor, at (C) inFIG. 2, will start building up with a gradient as at t₃. Then at t₄,when the capacitor voltage V_(c1) reaches the threshold V_(th1), the NOTcircuit 10 will go low as at (D) in FIG. 2.

The active switch Q₁ is shown turned off at t₅ in FIG. 2 whereupon theenergy that has been stored on the transformer 6 will be released. Thetransformer secondary N₂ will have induced thereon a voltage oriented ina direction opposite to that when the active switch Q₁ was “on.”Thereupon, as indicated by the arrow in FIG. 1 and diagramed at (F) inFIG. 2, the current I_(s) will flow along the path comprising thetransformer secondary N₂, smoothing capacitor C_(o), and synchronousrectifier Q₂. Containing the diode D_(o), the synchronous rectifier Q₂permits the flow of the current I_(s) therethrough irrespective ofwhether the switch 8 is on or off.

As indicated at (B) in FIG. 2, the drain-source voltage V_(ds) of thesynchronous rectifier Q₂ becomes zero or nearly so when the activeswitch Q₁ is turned off at t₅. Both inputs to the NOR circuit 12 of thesynchronous rectifier circuit 7 become low at that moment, so that itwill go high as at (E) in FIG. 2. Applied between the gate G and sourceS of the synchronous rectifier circuit 2, this high output from the NORcircuit 12 will cause conduction through the synchronous rectifierswitch 8. The current will then flow from source S toward drain D of thesynchronous rectifier Q₂.

The charging of the capacitor C₁ of the synchronous rectifier controlcircuit 7 will terminate when the active switch Q₁ is turned off as att₅ because then the drain-source voltage V_(ds) of the synchronousrectifier Q₂ becomes zero or nearly so as aforesaid. Then the capacitorC₁ will start discharging through the resistor R₂ of thecharge/discharge circuit 9 with a predetermined time constant. At (C) inFIG. 2 is shown the capacitor voltage V_(c1) as diminishing as from t₅to t₇.

Then, at t₆, when the capacitor voltage V_(c1) drops to the thresholdV_(th1), the NOT circuit 10 will go high as at (D) in FIG. 2. The NORcircuit 12 will go low, as at (E) in FIG. 2, in response to the highoutput from the NOT circuit 10, thereby causing nonconduction throughthe synchronous rectifier switch 8. The current I_(s) will continueflowing through the diode D_(o) of the synchronous rectifier Q₂ afterthe switch 8 has turned off. As will be understood from (F) in FIG. 2,the current I_(s) will flow only through the diode D_(o) for such abrief length of time, and will be of such small magnitude, that inpractice there will be no risk of the diode being ruptured.

The advantages gained by the foregoing embodiment of the invention maybe recapitulated as follows:

1. The conducting periods of the synchronous rectifier switch 8 isautomatically timed to the nonconducting periods of the active switch Q₁by the synchronous rectifier control circuit 7 of simplifiedconfiguration including the logic network 20, so that the synchronousrectifier switch is never to turn on during the conducting periods ofthe active switch.

2. The length of the conducting periods of the synchronous rectifierswitch 8 is accurately and easily adjustable by the charge and dischargetime constants of the capacitor C₁ of the synchronous rectifier circuit7 and the threshold value of the NOT circuit 10.

The first recited advantage will be better appreciated by consideringthe case where the synchronous rectifier switch 8 is set into conductionwhile the active switch Q₁ is conducting. Then, if the synchronousrectifier switch 8 permits bidirectional current flow therethrough, thecurrent will flow from transformer secondary N₂ to synchronous rectifierswitch 8, preventing proper rectification and possibly causing noiseproduction and circuit destruction. No such troubles will occur in theFIG. 1 apparatus as the synchronous rectifier switch 8 conducts onlyduring the nonconduction of the active switch Q₁.

Embodiment of FIG. 3

The second preferred form of switching-mode power supply shown in FIG. 3incorporates a modified synchronous rectifier circuit 3 _(a) and isidentical with the FIG. 1 embodiment in all the other details ofconstruction. The modified synchronous rectifier 3 _(a) differs from itsFIG. 1 counterpart 3 only in its synchronous rectifier control circuit 7_(a), which in turn differs from its FIG. 1 counterpart 7 only in itscharge/discharge circuit 9 _(a).

The modified charge/discharge circuit 9 _(a) features a dischargecircuit of a resistor R₂′ and diode D₂ interconnected in series, insteadof the discharge resistor R₂ of the FIG. 1 charge/discharge circuit 9.This discharge circuit is connected in parallel with the charge circuitcomprised of the diode D₁ and resistor R₁ which are interconnected inseries as in the FIG. 1 embodiment. The charge diode D₁ and dischargediode D₂ are opposite in polarity.

The second preferred form of switching-mode power supply operates likethat of FIG. 1 except for the path of the current discharged from thecapacitor C₁ of the synchronous rectifier control circuit 7 _(a), sothat the waveform timing diagram of FIG. 2 applies to this embodiment aswell. The current discharged from the capacitor C₁ during the periods oft₁–t₃, t₅–t₇, etc., flows along the path comprising the dischargeresistor R₂′, discharge diode D₂, and synchronous rectifier switch 8.

Embodiment of FIG. 4

This third preferred form of switching-mode power supply featuresanother modified synchronous rectifier control circuit 7 _(b) and anisolating transformer 23. These two components together with thesynchronous rectifier Q₂ constitute in combination a modifiedsynchronous rectifier circuit 3 _(b). All the other details ofconstruction are as previously described with reference to FIG. 1.

The synchronous rectifier control circuit 7 _(b) has an input connectedby way of a conductor 21 _(a) to the output conductor 16 of the activeswitch control circuit 5, and another input connected by way of aconductor 22 _(a) to the grounded DC input terminal 1 _(b). Thus thesynchronous rectifier control circuit 7 _(b) inputs thepulse-width-modulated switch control signal V_(g), shown at (A) in FIG.2, which is delivered from the switch control circuit 5 to the activeswitch Q₁ thereby to drive the same so as to keep the DC output voltageconstant. Instead, unlike its FIG. 1 counterpart 7, the synchronousrectifier control circuit 7 _(b) has no conductors 21 and 22 connectingthe same to the drain and source of the synchronous rectifier circuitQ₂.

Internally, the synchronous rectifier control circuit 7 _(b) isconfigured just like its FIG. 1 counterpart 7. The capacitor C₁ fordetermination of the conducting periods of the synchronous rectifierswitch 8 is connected between the pair of input conductors 21 _(a) and22 _(a). The NOR circuit 12 has one input connected by way of aconductor 11 _(a) to the input conductor 21 _(a) for inputting theswitch control signal V_(g), and another input to the NOT circuit 10.

The isolating transformer 23 has a primary winding 24 connected betweenthe output of the NOR circuit 12 and the grounded conductor 22 _(a). Itis understood that the unshown power supply for the NOT circuit 10 andNOR circuit 12 provides a preassigned voltage based upon the groundedconductor 22 _(a). The secondary winding 25 of the isolating transformer23 is connected between the gate and source of the synchronous rectifierQ₂.

The synchronous rectifier control circuit 7 _(b) operates like its FIG.1 counterpart 7 except that the capacitor C₁ is charged as dictated bythe switch control signal V_(g) being impressed to the active switch Q₁.The operation of the complete apparatus is therefore consideredself-evident from FIG. 2.

Embodiment of FIG. 5

The fourth preferred form of switching-mode power supply is akin to theFIG. 1 embodiment except for the addition of circuit means for limitingthe conducting periods of the synchronous rectifier switch 8. Theconduction limiter circuit, as it might be so called, includes a secondcapacitor C₂ (first capacitor being seen at C₁ in the synchronousrectifier control circuit 7) and a charge/discharge circuit 30 therefor.The charge/discharge circuit 30 comprises a charge resistor R₅ and adischarge diode D₃. The charge resistor R₅ is connected between thepositive DC output terminal 4 _(a) and one terminal of the secondcapacitor C₂, the other terminal of which is connected to the negativeDC output terminal 4 _(b) by way of the conductor 22. The dischargediode D₃ has its anode connected to the capacitor C₂ and its cathode tothe conductor 21.

Also included in the conduction limiter circuit is a NOT circuit 31which has its input connected to the second capacitor C₂ for shaping thevoltage across the second capacitor into a binary conduction limiteroutput signal. The output of the NOT circuit 31 is connected via a diode32 to the first capacitor C₁ of the synchronous rectifier controlcircuit 7. The diode 32 has its anode connected to the first capacitorC₁, and its cathode to the NOT circuit 31, for providing a compulsorydischarge path for the capacitor.

The FIG. 5 power supply is similar in operation to that of FIG. 1 exceptfor its conduction limiter circuit. Reference may be had to FIG. 6 forthe following operational description of the conduction limiter circuit.This figure indicates at (A) the switch control signal V_(g) for theactive switch Q₁, at (B) the voltage V_(c1) across the first capacitorC₁ for determination of the conducting periods of the synchronousrectifier switch 8, at (C) the output from the NOT circuit 10 of thesynchronous rectifier control circuit 7, at (D) the output from the NORcircuit 12 of the synchronous rectifier control circuit 7, at (E) thevoltage V_(c2) across the second capacitor C₂ of the conduction limitercircuit, and at (F) the output from the NOT circuit 31 of the conductionlimiter circuit.

The waveforms of FIG. 6 are plotted on the assumption that the powersupply had been operating normally until t₀ or t₁ and that then theconducting period of the synchronous rectifier switch 8 became longerthan normal. The capacitor C₂ of the conduction limiter circuit ischarged during the conducting periods T_(on) of the active switch Q₁ byway of the path comprising smoothing capacitor C_(o) and chargingresistor R₅. During the nonconducting periods T_(off) of the activeswitch Q₁, on the other hand, the capacitor C₂ discharges by way of thepath comprising the discharge diode D₃ and synchronous rectifier switch8, the path comprising the diode D₃ and the diode D₁ and resistors R₁and R₂ of the charge/discharge circuit 9, and the path comprising thedischarge diode D₃ and the resistors R₃ and R₄ of the synchronousrectifier control circuit 7. A rapid discharge will occur, as at (E) inFIG. 6, so small being the discharge time constant of the pathcomprising the discharge diode D₃ and synchronous rectifier switch 8.

As will be understood from (E) in FIG. 6, the capacitor C₂ of theconduction limiter circuit is so made that the voltage V_(c2) across thesame never builds up to the threshold V_(th2) of the NOT circuit 31 aslong as the power supply is operating normally. The NOT circuit 31 hastherefore been high until t₁, as at (F) in FIG. 6. The diode 32 has beenreverse biased, blocking the discharge path of the capacitor C₁. Theoperation of the complete power supply to this moment is as describedabove with reference to FIG. 2.

In the event of an abnormal drop in the DC output voltage due to asudden rise in the voltage requirement of the load 4, the active switchQ₁ will stay conductive, as at (A) in FIG. 6, even after t₁ when thevoltage V_(c2) across the second capacitor C₂ grows higher than thethreshold V_(th2) of the NOT circuit 31 as at (E) in FIG. 6. Thereuponthe NOT circuit 31 will go low as at (F) in FIG. 6. The diode 32 willthen be forward biased and so become conductive, thereby creating acompulsory discharge path comprising the first capacitor C₁, diode 32,and NOT circuit 31. The voltage V_(c1) across the first capacitor C₁will start diminishing at t₁, as at (B) in FIG. 6. Then, at t₂, when thefirst capacitor voltage V_(c1) drops below the threshold V_(th1) of theNOT circuit 10, this NOT circuit will go high as at (C) in FIG. 6. Thusdisabled, the NOR circuit 12 will be prevented from causing conductionthrough the synchronous rectifier switch 8.

The active switch Q₁ is shown to turn off at t₃ and remain so until t₄.During this t₃–t₄ nonconducting period T_(off) the transformer 6 willrelease its energy from its secondary N₂ along the path comprising thesmoothing capacitor C_(o) and the diode D_(o) of the synchronousrectifier Q₂. The synchronous rectifier switch 8 is also off during thet₃–t₄ period, performing no synchronous rectification. However, thisperiod of no synchronous rectification is so brief compared to that ofsynchronous rectification that the synchronous rectifier Q₂ willexperience no such temperature rise as to incur its own destruction. TheNOT circuit 31 goes high at t₃, as at (F) in FIG. 6, when the secondcapacitor C₂ discharges as at (E) in FIG. 6.

Let us assume that the capacitor C₁ of the synchronous rectifier controlcircuit 7 was not compulsorily discharged in response to the output fromthe NOT circuit 31. Then, as indicated by the dashed line at (B) in FIG.6, the voltage V_(c1) across this capacitor C₁ would continue buildingup after t₁, until t₃ when the active switch Q₁ went out of conduction.Then the capacitor voltage V_(c1) would decline from t₃ to t₄, duringwhich period the active switch Q₁ was off. As indicated also by thedashed line at (C) in FIG. 6, the NOT circuit 10 would stay low as longas the capacitor voltage V_(c1) was above its threshold V_(th1).

At t₃, when the active switch Q₁ turned off, the diode D_(o) of thesynchronous rectifier Q₂ would turn on, so that the conduction perioddetector circuit 11 of the synchronous rectifier control circuit 7 wouldgo low. Then the NOR circuit 12 would go high at t₃, as indicated by thedashed line at (D) in FIG. 6, thereby causing conduction through thesynchronous rectifier switch 8.

If the duration of the ensuing t₃–t₄ nonconducting period T_(off) isshot, the voltage V_(c1) across the capacitor C₁ of the synchronousrectifier control circuit 7 would be kept higher than the thresholdV_(th1) of the NOT circuit 10. The NOR circuit 12 would therefore remainhigh after t₄ when the active switch Q₁ was rendered conductive. Bothactive switch Q₁ and synchronous rectifier switch 8 would then beconductive after t₄. This certainly is a grave malfunctioning of theswitching-mode power supply, giving rise to noise problems and possiblerupture of the associated circuit elements.

By contrast, in this FIG. 5 embodiment, the synchronous rectifier switch8 does not turn on at t₃ and remains off even after t₄ when the activeswitch Q₁ turns on, all as indicated by the solid lines in FIG. 6.Concurrent closure of both active switch Q₁ and synchronous rectifierswitch 8 is thus avoided. The conduction limiter circuit introduced inthis embodiment is applicable to all the other embodiments disclosedherein.

Embodiment of FIG. 7

A voltage-regulating zener diode 33 is newly incorporated in this FIG. 7embodiment, which is otherwise identical in construction with that ofFIG. 5. The zener diode 33 is connected between the capacitor C₁ of thesynchronous rectifier control circuit 7 and the diode 32 of theconduction limiter circuit. Oriented opposite to the diode 32, the zenerdiode conducts when the voltage V_(c1) develops across the capacitor C₁above a prescribed limit.

The operation of this power supply will become apparent from aconsideration of FIG. 8, which shows the waveforms appearing at the sameparts of the FIG. 7 embodiment as those of the FIG. 5 embodiment atwhich appear the waveforms given in FIG. 6. A comparison of FIGS. 6 and8 will further reveal that the waveforms (A), (E) and (F) are the samein both figures.

When the NOT circuit 31 of the conduction limiter circuit goes low att₁, as at (F) in FIG. 8, the current discharged from the capacitor C₁ ofthe synchronous rectifier control circuit 7 will flow into the NOTcircuit 31 via the zener diode 33 and diode 32. As seen at (B) in FIG.8, the voltage V_(c1) across the capacitor C₁ will dwindle from t₁ tot₂, the latter being the moment the capacitor voltage becomes less thanthe sum of the zener voltage of the zener diode 33 and the voltageacross the diode 32. Then the zener diode 33 will become nonconductive,terminating the discharge of the capacitor C₁, with the result that thecapacitor voltage V_(c1) is maintained thereafter at the value at t₂which is equal to the zener voltage.

This value of the capacitor voltage V_(c1) is shown to be slightlyhigher than the threshold V_(th1) of the NOT circuit 10 of thesynchronous rectifier control circuit 7 at (B) in FIG. 8. Then thecapacitor voltage V_(c1) will start diminishing at t₃, when the activeswitch Q₁ goes off as at (A) in FIG. 8, and keep doing so until t₃′ whenthe capacitor voltage becomes less than the threshold V_(th1) of the NOTcircuit 10. Both inputs to the NOR circuit 12 of the synchronousrectifier control circuit 7 are therefore low from t₃ to t₃′, so thatits output is high as at (D) in FIG. 8, causing conduction through thesynchronous rectifier switch 8.

The synchronous rectifier switch 8 will go off at t₃′ when the capacitorvoltage V_(c1) becomes less than the threshold V_(th1) of the NOTcircuit 10 as above, because then the NOT circuit 10 will go high as at(C) in FIG. 8 thereby causing the NOR circuit 12 to go low.

The FIG. 7 embodiment offers the advantage, in addition to those setforth in conjunction with that of FIG. 1, that the high periods of theNOR circuit 12 are variable through adjustment of the zener voltage ofthe zener diode 33. An adjustment of the high periods of the NOR circuit12 is tantamount to that of the conducting periods of the synchronousrectifier switch 8.

It will be apparent that the serial circuit of the discharge resistorR₂′ and diode D₂, FIG. 3, is adoptable in substitution for the dischargeresistor R₂ in the embodiments of FIGS. 5 and 7. An equivalent of theconduction limiter circuit of FIG. 5 or 7 may be added to the FIG. 4embodiment as well.

Embodiment of FIG. 9

A wave-shaping circuit shown at 40 in FIG. 9 finds use in theswitching-mode power supply of FIG. 1 in place of the NOT circuit 10 ofthe synchronous rectifier control circuit 7. The wave-shaping circuit 40includes a comparator 41 having a negative input connected to thecapacitor C₁, FIG. 1, and a positive input connected to a referencevoltage source 42. The reference voltage source 42 provides a voltagefunctionally equivalent to the threshold voltage V_(th1) of the NOTcircuit 10. The output of the comparator 41 is coupled to one of theinputs of the NOR circuit 12, FIG. 1.

It is now apparent that just like the NOT circuit 10, the wave-shapingcircuit 40 functions to convert the voltage V_(c1) across the capacitorC₁ into a binary signal prior to application to the NOR circuit 12. Thiswave-shaping circuit may be employed in lieu of the NOT circuit 10 inFIGS. 3, 4, 5 and 7 and of the NOT circuit 31 in FIGS. 5 and 7.

Embodiment of FIG. 10

A synchronous rectifier switch driver circuit 50, FIG. 10, may beinserted between the NOR circuit 12 of the synchronous switch controlcircuit and the gate G of the synchronous rectifier switch 8, in any ofthe foregoing embodiments of the invention.

The switch driver circuit 50 includes an npn transistor 51 and a pnptransistor 52 which have their bases interconnected and furtherconnected to the NOR circuit 12 via a resistor 53. The transistor 51 hasits collector connected to the DC output terminal 4 _(a) via a resistor54 whereas the other transistor 52 has its collector connected to theother DC output terminal 4 _(b). The emitters of both transistors 51 and52 are interconnected and further connected to the gate G of thesynchronous rectifier Q₂ via a resistor 55. Another resistor 56 isconnected between the gate G and source S of the synchronous rectifierQ₂.

The transistor 51 conducts each time the NOR circuit 12 goes high,causing conduction through the synchronous rectifier switch 8. The sameswitch driver circuit could be added to any of the embodiments of FIGS.2, 5 and 7.

Embodiment of FIG. 11

This embodiment is similar in construction to that of FIG. 1 except fora modified synchronous rectifier control circuit 7 _(c) which has inputsconnected to the transformer primary N₁ and outputs connected to thesynchronous rectifier circuit 8 via an isolating transformer 23. Thesynchronous rectifier control circuit 7 _(c) is functionally equivalentto its FIG. 1 counterpart 7. The isolating transformer 23 is bothfunctionally and constructionally identical with that designated by thesame reference numeral in FIG. 4.

The modified synchronous rectifier control circuit 7 _(c) includes thecapacitor C₁ having one electrode connected to the drain of the activeswitch Q₁ via a charge/discharge circuit 9 _(b). The other electrode ofthe capacitor C₁ is connected to the source of the active switch Q₁.

Like its FIG. 3 counterpart 9 _(a), the charge/discharge circuit 9 _(b)comprises a serial connection of a charge diode D₁ and charge resistorR₁ and a serial connection of a discharge diode D₂ and dischargeresistor R₂′. The charge resistor R₁ has one extremity connected to thedrain of the active switch Q₁ via the charge diode D₁ and anotherextremity connected to the capacitor C₁. The discharge resistor R₂′ hasone extremity connected to the drain of the active switch Q₁ via thedischarge diode D₂ and another extremity connected to the capacitor C₁.Thus, via the charge/discharge circuit 9 _(b), the capacitor C₁ ischarged during the nonconducting periods of the active switch Q₁ anddischarges during the conducting periods of the active switch.

Another component of the modified synchronous rectifier control circuit7 _(c) is a NOR circuit 60 which is functionally akin to the logicnetwork 20 of the FIG. 1 embodiment. The NOR circuit 60 has one inputconnected via the conductor 11 _(a) to the conductor 21 _(a) forinputting the switch control signal V_(g) being applied from switchcontrol circuit 5 to active switch Q₁. Another input of the NOR circuit60 is connected to the capacitor C₁. Between the pair of supplyterminals of the NOR circuit 60 is connected a capacitor 62 by way of apower supply therefor. The capacitor 62 has one electrode connected viaa rectifier diode 61 to the conductor 21 _(a) and the other electrode tothe source of the active switch Q₁.

The NOR circuit 60 compares the incoming voltage across the capacitor C₁with a predetermined threshold V_(th2), FIG. 12 (C), and furthercompares the resulting binary signal with the switch control signalV_(g). As is apparent from (A), (C) and (D) in FIG. 12, the NOR circuit60 goes high for causing conduction through the synchronous rectifierswitch 8 when the voltage V_(c1) across the capacitor C₁ is less thanthe threshold V_(th2) and, at the same time, when the active switch Q₁is off.

A comparison of (A) and (C) in FIG. 12 will further reveal that thecapacitor C₁ is charged during the nonconducting periods T_(off) of theactive switch Q₁ and discharges during the conducting periods T_(on)thereof. This is a reversal of the case, indicated at (A) and (C) inFIG. 2, for the capacitor C₁ of the FIG. 1 embodiment.

The modified synchronous rectifier control circuit 7 _(c) is shown asfurther comprising a synchronous rectifier switch driver circuit 50_(a). Like its FIG. 10 counterpart 50 this driver circuit 50 _(a)comprises two transistors 51 and 52 and two resistors 53 and 54. Thetransistor 51 has its collector connected via the resistor 54 to asource 63 of a supply voltage V_(cc) whereas the other transistor 52 hasits collector connected to the grounded conductor 22 _(a). The emittersof both transistors 51 and 52 are both connected to a coupling capacitor64, thence to the isolating transformer 23, and thence to thesynchronous rectifier switch 8.

Additionally, in this FIG. 11 embodiment, the switch control circuit 5is connected to the active switch Q₁ via two resistors R₁₁ and R₁₂. Thetransformer 23 is likewise connected to the synchronous rectifier switch8 via two resistors R₁₃ and R₁₄.

The NOR circuit 60 of the FIG. 11 synchronous rectifier control circuit7 _(c) functions just like the logic network 20 of the FIG. 1 circuit 7.It is therefore apparent that this embodiment offers the same advantagesas does that of FIG. 1.

Embodiment of FIG. 13

A further modified synchronous rectifier control circuit 7 _(d) shown inFIG. 13 finds use in the FIG. 1 power supply in substitution for thecircuit 7. The modified synchronous rectifier control circuit 7 _(d)features another modification 9 _(c) of the charge/discharge circuit 9of the FIG. 1 embodiment. The modified charge/discharge circuit 9 _(c)has the resistor R₁ but neither discharge resistor R₂ nor rectifierdiode D₁ of the FIG. 1 circuit 9. Connected between conductor 21 andcapacitor C₁, the resistor R₁ lends itself to use for both charging anddischarging of the capacitor C₁.

Embodiment of FIG. 14

FIG. 14 shows a further modification 7 _(e) of the synchronous rectifiercontrol circuit 7 of FIG. 1. The modified synchronous rectifier controlcircuit 7 _(e) features a further modified charge/discharge circuit 9_(d) but is otherwise identical with its FIG. 1 counterpart 7. Themodified charge/discharge circuit 9 _(d) is similar in construction toits FIG. 1 counterpart 9 except for the absence of the diode D₁. Theresistor R₁, like that of FIG. 13, is utilized for both charging anddischarging the capacitor C₁.

Embodiment of FIG. 15

A further modified synchronous rectifier control circuit 7 _(f) given inFIG. 15 incorporates a modified logic network 20 _(a), the other detailsof construction being as above explained with reference to FIG. 1. Themodified logic network 20 _(a) features an inverting AND gate 12 _(a) inplace of the NOR circuit or OR-inverter combination 12 of FIG. 1. Theinverting AND gate 12 _(a) is functionally equivalent to the NOR circuit12.

Embodiment of FIG. 16

In FIG. 16 is shown a further modified synchronous rectifier controlcircuit 7 _(g) which is of the same construction as its FIG. 1counterpart 7 except for another modified logic network 20 _(b). Thelogic network 20 _(b) differs from that of FIG. 1 in having a two-inputNOR circuit 10 _(a) in substitution for the NOT circuit 10. The NORcircuit 10 _(a) has its two inputs both connected to the capacitor C₁and its output connected to the NOR circuit 12. Having its two inputsshort-circuited, the NOR circuit 10 _(a) is functionally equivalent tothe NOT circuit 10.

Embodiment of FIG. 17

Still another modified logic network 20, is included in the synchronousrectifier control circuit 7 _(h) of FIG. 17, which is otherwise of thesame construction as that of FIG. 1. The logic network 20 _(c) is acombination of an AND circuit 10 _(b) and NOT circuit 12 _(b). The ANDcircuit 10 _(b) has one input connected via the NOT circuit 12 _(b) tothe junction 13 between the voltage-dividing resistors R₃ and R₄. As hasbeen stated in connection with FIG. 1, the potential at the junction 13represents a predefined fraction of the voltage between the drain andsource of the synchronous rectifier Q₂. The other input of the ANDcircuit 10 _(b) is connected to the capacitor C₁.

It is understood that the AND circuit 10 _(b) is equipped to compare theincoming voltage across the capacitor C₁ with a predetermined threshold.Only when the capacitor voltage is higher than the threshold, and at thesame time when the output from the NOT circuit 12 _(b) is high, does theAND gate 10 _(b) go high thereby causing conduction through the switch8, FIG. 1, of the synchronous rectifier Q₂. The logic network 20 _(c) istherefore functionally equivalent to its FIG. 1 counterpart 20.

Embodiment of FIG. 18

FIG. 18 represents an application of the instant invention to theboost-converter variety of switching-mode power supply. This embodimentdiffers from that of FIG. 1 only in the design of its boost convertercircuit 2 _(a) and its connections to the other parts of the apparatus.

The boost converter circuit 2 _(a) includes an inductor 6 _(a) which isakin in construction to the transformer 6, FIG. 1, minus the secondarywinding N₂. Coiled around the magnetic core 6′, the winding N₁ of theinductor 6 _(a) is connected in series with the active switch Q₁. Theinductor 6 _(a) stores energy as does the transformer 6. It isunderstood that the inductor 6 _(a) has another winding, not shown,which is electromagnetically coupled to the winding N₁ for powering thevarious circuit elements. The DC output terminal 4 _(a) is connected tothe junction between inductor winding N₁ and active switch Q₁, and theother DC output terminal 4 _(b) to the DC input terminal 1 _(b) via thesynchronous rectifier Q₂.

Energy is stored on the inductor 6 _(a) during the conducting periods ofthe active switch Q₁. During its nonconducting periods the sum of thevoltages across the DC source 1 and across the inductor winding N₁causes the smoothing capacitor C_(o) to be charged to a voltage higherthan the DC input voltage. The synchronous rectifier Q₂ turns on and offin the same time relationship to the active switch Q₁ as in the FIG. 1embodiment.

Embodiment of FIG. 19

The invention is applicable to a forward-converter type ofswitching-mode power supply as well, as in FIG. 19. The forwardconverter circuit 2 _(b) of this embodiment differs from its FIG. 1counterpart 2 only in having a transformer 6 _(b) whose primary N₁ andsecondary N₂ are oriented in the same direction. As a consequence, thesynchronous rectifier Q₂ conducts during the conducting periods of theactive switch Q₁. The synchronous rectifier control circuit 7 _(i) ofthis embodiment is drawn in block form because it is of the sameconstruction as its FIG. 1 counterpart 7 except that, as in the FIG. 11embodiment, the equivalent of the capacitor C₁ is charged during thenonconducting periods of the active switch Q₁ and discharges during itsconducting periods.

This embodiment incorporates a smoothing circuit comprising an inductorL_(o) and a second synchronous rectifier Q₃, in addition to thesmoothing capacitor C_(o). The inductor L_(o) is connected betweentransformer secondary N₂ and smoothing capacitor C_(o). The secondsynchronous rectifier Q₃ is connected in parallel with the serialcircuit of the inductor L_(o) and smoothing capacitor C_(o). The energythat has been stored on the inductor L₀ during the conducting periods ofthe active switch Q₁ is released by way of the path comprising theinductor L_(o), smoothing capacitor C_(o), and second synchronousrectifier Q₃.

Fabricated in the form of a field-effect transistor, the secondsynchronous rectifier Q₃ comprises a parallel connection of switch 70and diode 71. The diode 71 may be either an integral part of the switch70 or a discrete part.

A control circuit 7 _(i)′ for the second synchronous rectifier Q₃ isalso shown in block form because it is of the same construction as thesynchronous rectifier control circuit 7 of FIG. 1. The secondsynchronous rectifier control circuit 7 _(i)′ operates to causeconduction through the second synchronous rectifier control switch 70when the second synchronous rectifier Q₃ must be conducting. Power lossat the second synchronous rectifier Q₃ is thus lessened.

The two synchronous rectifier control circuit 7 _(i) and 7 _(i)′ used inthis embodiment are each constructed like the FIG. 1 circuit 7, so thatthis embodiment obtains the same advantages as those of the FIG. 1embodiment.

Embodiment of FIG. 20

In FIG. 22 is shown the invention as applied to a chopper-typeswitching-mode power supply having a converter circuit 2 _(c) whichcontains an active switch Q₁ in the form of a pnp transistor. The activeswitch Q₁ has an emitter connected to one DC input terminal 1 _(a), acollector connected to one DC output terminal 4 _(a) via the inductorL_(o), and a base connected to the other DC input terminal 1 _(b) via aswitch control circuit 5′ of prior art design by which the switch isturned on and off.

Connected between the active switch 1 and the pair of DC outputterminals 4 _(a) and 4 _(b) is a smoothing circuit of the sameconstruction as in FIG. 19, comprising the smoothing capacitor C_(o),the inductor L_(o) and the synchronous rectifier Q₃. The synchronousrectifier Q₃ and its control circuit 7 _(i)′ are similar in function tothose designated by the same reference characters in FIG. 19 and inconstruction to the synchronous rectifier Q₂, FIG. 1, and its controlcircuit 7.

Embodiment of FIG. 21

The switching-mode power supply shown here is broadly divisible into aninverting push-pull converter circuit 2 _(d) and a rectifying andsmoothing circuit. Itself well known in the art, the inverting push-pullconverter circuit 2 _(d) comprises a pair of active switches Q₁₁ and Q₁₂and a transformer 6 _(d). The active switches Q₁₁ and Q₁₂, each in theform of a transistor, are to be alternately turned on and off by aswitch control circuit 5″ which has outputs connected to their bases.

The transformer 6 _(d) has both of its primary winding N₁ and secondarywinding N₂ center-tapped. The transformer primary N₁ has its tapconnected to one DC input terminal 1 _(a), its one extremity connectedto the other DC input terminal 1 _(b) via the active switch Q₁₁, and itsother extremity connected to this other DC input terminal 1 _(b) via theother active switch Q₁₂. The transformer secondary N₂ is connected tothe pair of DC output terminals 4 _(a) and 4 _(b) via a full-waverectifier circuit and the smoothing capacitor C_(o).

The full-wave rectifier circuit comprises a first synchronous rectifierQ₂ connected between one extremity of the transformer secondary N₂ andone DC output terminal 4 _(a), and a second synchronous rectifier Q₂′connected between the other extremity of the transformer secondary N₂and that one DC output terminal 4 _(a). The center tap of thetransformer secondary N₂ is connected to the other DC output terminal 4_(b). The two synchronous rectifiers Q₂ and Q₂′ are both of the sameconstruction as its FIG. 1 counterpart Q₂.

For on/off control of the synchronous rectifiers Q₂ and Q₂′ there areprovided two synchronous rectifier control circuits 7 _(j) and 7 _(j)′for the respective rectifiers. The synchronous rectifier controlcircuits 7 _(j) and 7 _(j)′ are each analogous in construction with theFIG. 1 circuit 7 except that the circuits 7 _(j) and 7 _(j)′ are so madethat the first synchronous rectifier Q₂ conducts during the conductingperiod of the first active switch Q₁₁, and that the second synchronousrectifier Q₂′ conducts during the conducting periods of the secondactive switch Q₁₂. The other constructional and operational details ofthis embodiment, as well as the advantages accruing therefrom, are aspreviously explained with reference to FIGS. 1 and 2.

POSSIBLE MODIFICATIONS

Although the switching-mode power supply according to the presentinvention has been shown and described hereinbefore in terms of somecurrently preferred forms, it is not desired that the invention belimited by the exact details of these preferred forms or by thedescription thereof. The following is a brief list of possiblemodifications of the illustrated embodiments which are all believed tofall within the purview of the instant invention:

1. The converter circuits 2 and 2 _(a)–2 _(d) of the FIGS. 1 and 18–21embodiments are replaceable as by half-bridge inverters orpolarity-inverting DC-to-DC converters.

2. The synchronous rectifier control circuits of the FIGS. 3–5, 7, 9–11and 13–17 embodiments are adoptable in places of the synchronousrectifier control circuits 7 _(i), 7 _(i)′, 7 _(j) and 7 _(j)′ of theFIGS. 18–21 embodiments.

3. The synchronous rectifier control circuits 7 _(d)–7 _(h) of the FIGS.13–17 embodiments are adoptable, either with or without minormodifications, in places of their counterparts in the FIGS. 4, 5, 7 and9–11 embodiments.

4. The FIGS. 3–5 and 7 embodiments are adaptable for a boost convertertype like the FIG. 11 embodiment.

5. The synchronous rectifier Q₂ could be serially connected between thetransformer secondary N₂ or primary N₁ and the positive supply terminal4 _(a).

6. The active switch Q₁ of the FIGS. 1, 3–7, 11, 18 and 19 could besemiconductor switches other than IGFETs, examples being transistors andinsulated-gate bipolar transistors (IGBTs).

7. The active switches Q₁, Q₁₁ and Q₁₂ of the FIGS. 20 and 21embodiments could also be of other types such as FETs and IGBTs.

8. The synchronous rectifiers Q₂ and Q₂′ could be of combinations ofother types of semiconductor switches such as a transistor or IGBT, anda diode.

1. A switching-mode power supply for DC-to-DC conversion, comprising:(a) DC input means for inputting a DC voltage; (b) DC output means foroutputting a DC voltage; (c) a converter circuit connected between theDC input means and the DC output means, the converter circuit includingan active switch for switching the input DC voltage on and off; (d) aswitch control circuit connected to the active switch of the convertercircuit for on/off control thereof; (e) a synchronous rectifier switchconnected between the converter circuit and the DC output means; (f) adiode connected in parallel with the synchronous rectifier switch; (g)conduction period detect means for providing a conduction period detectsignal indicative of periods of time during which the active switch isheld conductive by the switch control circuit; (h) a capacitor fordetermination of conducting periods of the synchronous rectifier switch;(i) a charge/discharge circuit connected to the converter circuit andthe capacitor for causing the latter to be charged and dischargedaccording to whether the active switch is conductive or nonconductive;and (j) a logic network having an input connected to the conductionperiod detect means, another input connected to the capacitor, and anoutput connected to the synchronous rectifier switch, for making on/offcontrol of the synchronous rectifier switch according to whether theactive switch is conducting or nonconducting and whether the voltageacross the capacitor is higher than a predefined voltage or not.
 2. Aswitching-mode power supply as defined in claim 1, wherein theconduction period detect means is adapted to provide a binary conductionperiod detect signal indicative of whether the active switch isconductive or nonconductive.
 3. A switching-mode power supply as definedin claim 1, wherein the conduction period detect means comprises meansfor detecting a voltage indicative of whether the active switch isconductive or nonconductive.
 4. A switching-mode power supply as definedin claim 1, wherein the charge-discharge circuit is adapted to cause thecapacitor to be charged during the conducting periods of the activeswitch and to discharge during the nonconducting periods of the activeswitch.
 5. A switching-mode power supply as defined in claim 1, whereinthe charge-discharge circuit comprises a resistor connected between oneterminal of the synchronous rectifier switch and one terminal of thecapacitor, the capacitor having another terminal connected to anotherterminal of the synchronous rectifier switch.
 6. A switching-mode powersupply as defined in claim 5, wherein the charge-discharge circuitcomprises a second resistor connected in parallel with the capacitor. 7.A switching-mode power supply as defined in claim 6, wherein thecharge-discharge circuit comprises a diode connected in series with thefirst recited resistor.
 8. A switching-mode power supply as defined inclaim 1 wherein the charge-discharge circuit comprises: (a) chargecircuit means comprising a first diode and a first resistor which areserially interconnected between one terminal of the synchronousrectifier switch and one terminal of the capacitor, the capacitor havinganother terminal connected to another terminal of the synchronousrectifier switch; and (b) discharge circuit means comprising a seconddiode and a second resistor which are serially interconnected betweensaid one terminal of the capacitor and said one terminal of thesynchronous rectifier switch.
 9. A switching-mode power supply asdefined in claim 1, wherein the DC input means comprises a pair of DCinput terminals, wherein the converter circuit comprises an inductor,wherein the active switch has a first main electrode connected to one ofthe DC input terminals via the inductor, a second main electrodeconnected to the another of the DC input terminals and a controlelectrode connected to the switch control circuit, wherein theconduction period detect means comprises a conduction period detectconductor connected to the switch control circuit for detecting a switchcontrol signal being thereby applied to the active switch for on/offcontrol thereof, wherein the charge/discharge circuit comprises a chargecircuit having a rectifying diode and a resistor which areinterconnected in series between the conduction period detect conductorand one terminal of the capacitor, wherein the charge/discharge circuitfurther comprises a discharge circuit connected in parallel with thecapacitor, and wherein the capacitor has another terminal connected tothe second main electrode of the active switch.
 10. A switching-modepower supply as defined in claim 1, wherein the DC input means comprisesa pair of DC input terminals, wherein the converter circuit comprises aninductor, wherein the active switch has a first main electrode connectedto one of the DC input terminals via the inductor, a second mainelectrode connected to the another of the DC input terminals and acontrol electrode connected to the switch control circuit, wherein theconduction period detect means comprises a conduction period detectconductor connected to the switch control circuit for detecting a switchcontrol signal being thereby applied to the active switch for on/offcontrol thereof, wherein the charge/discharge circuit comprises a chargecircuit having a first rectifying diode and a resistor which areinterconnected in series between the first main electrode of the activeswitch and the capacitor, and wherein the charge/discharge circuitfurther comprises a discharge circuit having a second rectifying diodeand a resistor which are interconnected in series between the capacitorand the first main electrode of the active switch.
 11. A switching-modepower supply as defined in claim 1, further comprising isolation meansconnected between the logic network and the synchronous rectifierswitch.
 12. A switching-mode power supply as defined in claim 1, whereinthe logic network comprises: (a) first logic circuit means having aninput connected to the capacitor for providing a binary capacitorvoltage signal indicative of results of comparison of the voltage acrossthe same with the predefined voltage; and (b) second logic circuit meanshaving an input connected to the conduction period detect means forinputting the conduction period detect signal, another input connectedto the first logic circuit means for inputting the capacitor voltagesignal, and an output connected to the synchronous rectifier switch, forcausing conduction through the synchronous rectifier switch when thecapacitor voltage is higher than the predefined voltage and at the sametime when the active switch is nonconducting.
 13. A switching-mode powersupply as defined in claim 12, wherein the first logic circuit meanscomprises a NOT circuit having a threshold value.
 14. A switching-modepower supply as defined in claim 12, wherein the first logic circuitmeans comprises a NOR circuit having two inputs both connected to thecapacitor.
 15. A switching-mode power supply as defined in claim 12,wherein the first logic circuit means of the logic network comprises acomparator having a first input connected to the capacitor and a secondinput connected to a reference voltage source.
 16. A switching-modepower supply as defined in claim 12, wherein the second logic circuitmeans of the logic network comprises a NOR or inverting AND circuit. 17.A switching-mode power supply as defined in claim 1, wherein the logicnetwork comprises: (a) a NOT circuit having an input connected to theconduction period detect means; and (b) an AND circuit having an inputconnected to the capacitor, another input connected to the NOT circuit,and an output connected to the synchronous rectifier switch, the ANDcircuit having a threshold value with which to compare the voltageacross the capacitor.
 18. A switching-mode power supply as defined inclaim 1, further comprising a conduction period limiter circuit forlimiting the conducting periods of the synchronous rectifier switch, theconduction limiter circuit comprising: (a) a second capacitor; (b) asecond charge/discharge circuit connected to the second capacitor forcausing the same to be charged during the conducting periods of theactive switch and to discharge during the nonconducting periods thereof;(c) a wave-shaping circuit connected to the second capacitor forproviding an output indicative of whether a voltage across the secondcapacitor is higher than a second predefined voltage or not, the secondpredefined voltage being such that the output from the wave-shapingcircuit indicates whether the conducting periods of the active switch isless than a prescribed limit or not; and (d) compulsory discharge pathmeans connected between the wave-shaping circuit and the first recitedcapacitor for providing a compulsory discharge path for the firstcapacitor.
 19. A switching-mode power supply as defined in claim 18,wherein the compulsory discharge path means of the conduction limitercircuit comprises a second diode so oriented as to conduct when theactive switch remains conductive longer than the prescribed limit.
 20. Aswitching-mode power supply as defined in claim 19, wherein thecompulsory discharge path means of the conduction limiter circuitfurther comprises a voltage-regulating diode connected in series with,and oriented opposite to, the second diode.
 21. A switching-mode powersupply as defined in claim 1, wherein the diode is built into thesynchronous rectifier switch.